View MIC913106_4197628.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

MIC9131
Micrel, Inc.
MIC9131
High-Voltage, High-Speed Telecom DC-to-DC Controller
General Description
The MIC9131 is a current-mode PWM controller that efficiently converts -48V telecom voltages to logic levels. The MIC9131 features a high voltage start-up circuit that allows the device to be connected to input voltages as high as 180V. The high input voltage capability protects the MIC9131 from line transients that are common in telecom systems. The start-up circuitry also saves valuable board space and simplifies designs by integrating several external components. The MIC9131 is capable of high speed operation. Typically the MIC9131 can control a sub-25ns pulse width on the gate out pin. Its internal oscillator can operate over 2.5MHz, with even higher frequencies available through synchronisation. The high speed operation of the MIC9131 is made safe by the very fast, 34ns response from current sense to output, minimizing power dissipation in a fault condition. The MIC9131 allows for the designs of high efficiency power supplies. It can achieve efficiencies over 90% at high output currents. Its low 1.3mA quiescent current allows high efficiency even at light loads. The MIC9131 has a maximum duty cycle of 75%. For designs requiring a maximum duty cycle of 50%, refer to the MIC9130. The MIC9131 is available in a 16-pin SOP and 16-pin QSOP package options. The junction temperature range is from -40C to +125C.
Features
* * * * * * * * * * * * * * * * * * * * * * Input voltages up to 180V Internal oscillator capable of > 2.5MHz operation Accurate 75% maximum duty cycle Synchronisation capability to 6MHz Current sense delay of 34ns Minimum pulse width of <25ns 90% efficiency 1.3mA quiescent current 1A shutdown current Soft-start Resistor programmable current sense threshold Selectable soft-start retry 4 sink, 12 source output driver Programmable under-voltage lockout Constant-frequency PWM current-mode control 16-pin SOIC and 16-pin QSOP Telecom power supplies Line cards ISDN network terminators Micro- and pico-cell base stations Low power (< 100W) dc-dc converters DSL line cards
Applications
Typical Application
VIN 36V to 72V
1M
0.1F
12V
10 10F 25V
1N5818
4:1 6:1 L = 530H
VOUT 3.3V @ 20A
Si4884DY (x2) B320A
1.5H 100F
37.4k
13
1
2
10
UVLO
LINE
VCC
7
FB COMP RBIAS SS CPWR VBIAS MIC9131
EN
Si4884DY (x2)
10k
6 3 12 8
OUT
16
SLOPE COMPENSATION
IRFS3IN20D
ISNS
14
8.2k
562k 0.1F
9
OSC SYNC AGND PGND
4
5
0.1 1W
11
15
10nF
2.61k 270pF
OPTO FEEDBACK
90% Efficient Telecommunications Power Supply
Micrel, Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel + 1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
August 2006
1
M9999-080206
MIC9131
Micrel, Inc. Part Number Max. Duty Cycle 75% 75% Junction Temp. Range -40C to +125C -40C to +125C Package 16-Pin SOP 16-Pin QSOP
Ordering Information
Standard MIC9131BM MIC9131BQS Pb-Free MIC9131YM MIC9131YQS
Pin Configuration
LINE 1
VCC 2
16 OUT 15 PGND 14 ISNS 13 UVLO 12 SS 11 AGND
RBIAS 3
OSC 4
SYNC 5
COMP 6
FB 7
10 EN 9 VBIAS
CPWR 8
16-Pin SOP (M) 16-Pin QSOP (QS)
Pin Description
Pin Number 1 2 3 4 5 6 7 8 Pin Name LINE VCC RBIAS OSC SYNC COMP FB CPWR Pin Function Line (Input): 180Vdc maximum supply input. May be floated if unused. Supply (Input): MIC9131 internal supply input. Bias Resistor (External Component): Connect 562K to ground. Oscillator RC Network (External Components): Connect external resistor-capacitor network to set oscillator frequency. Synchronization (Input): External oscillator input for slave operation of controller. See OSC. Do not float. Compensation (External Components): Error amplifier output for external compensation network connection. Feedback (Input): Error amplifier inverting input. Current Limit Selection (Input): When CPWR is high, an over-current condition at the ISNS input will terminate the gate drive and reset the soft-start latch. If the CPWR pin is low, an over-current condition at the ISNS input will terminate the gate drive signal, but will not cause a reset of the soft-start circuit. Reference (Output): Internal 5V supply. Will source 5mA maximum. Enable (Input): Logic level enable/shutdown input; logic high = enabled (on), logic low = shutdown (off). Analog Ground (Return) Soft-Start (External Components): Connect external capacitor to slowly ramp up duty cycle during startup and over-current conditions. Undervoltage Lockout (External Components): Connect to unbiased resistive divider network to set controller's minimum operating voltage. Connect to VBIAS if not needed. Current Sense (Input): Connect between external switching MOSFET source and switch current sense resistor. Power Ground (Return) Switch Drive Output (Output): Connect to gate of external switching MOSFET.
9 10 11 12 13
VBIAS EN AGND SS UVLO
14 15 16
ISNS PGND OUT
M9999-080206
2
August 2006
MIC9131
Micrel, Inc.
Absolute Maximum Ratings (Note 1)
Line Input Voltage (VLINE) ......................................... +190V VCC Input Voltage (VCC) .............................................. +19V Current Sense Input Voltage (VISNS) .............. -0.3 to +5.3V Enable Voltage (VEN) ............................ -0.3 to VCC + 0.3V Feedback Input Voltage (VFB) ........................ -0.3 to +5.3V Sync Input Voltage (VSYNC) ............................ -0.3 to +5.3V Soft-Start Voltage (VSS) .................................. -0.3 to +5.3V UVLO Voltage (VUVLO) ................................... -0.3 to +5.3V Storage Temperature (TS) ........................ -65C to +150C Power Dissipation (PD) ........................................................ 16-pin SOP ...................................400mW @ TA = +85C 16-pin QSOP ................................245mW @ TA = +85C ESD Rating, Note 3
Operating Ratings (Note 2)
Line Input Voltage (VLINE) .................VCC to +180V, Note 4 VCC Input Voltage (VCC) .................................. +9V to +18V Junction Temperature Range (TJ) ............ -40C to +125C Package Thermal Resistance 16-pin SOP (JA) .............................................. 100C/W 16-pin QSOP (JA) ............................................ 163C/W
Electrical Characteristics
TA = 25C, VLINE = 48V, VCC = 10V, Rt = 9.47K, Ct = 470pF, RBIAS = 562k, VEN = 10V, VISNS = 0V, VUVLO = 2V, VSYNC = 0V, unless otherwise noted. Bold values indicate -40 C TJ +125C. Parameter Condition Min 4.7 4.6 24 5 180 200 fOSC/4 75 2.5 100 6 2.5 0.7 50 VCOMP = VFB 2.475 2.45 2.5 90 4 9V VCC 18V 60 80 1 3.5 100 2.5 115 4 160 1.5 1.5 300 VFB = 2.7V; VCOMP = 5V VFB = 2.7V; ICOMP = -50A VFB = VCOMP VFB = 2.3V; VCOMP = 0V 2.525 2.55 MHz V V ns V dB MHz dB A mA mV V nA V/s V/s Typ Max 5.0 5.1 40 30 220 Bias Regulator Output Voltage Line Regulation Load Regulation Oscillator Section Initial Accuracy (fOSC) Maximum Duty Cycle Voltage Stability (f/f) Temperature Stability ppm/C Maximum Sync Frequency Sync Threshold Level Sync Hysteresis Sync Minimum Pulse Width Error Amp Section FB Voltage Open Loop Voltage Gain, AVOL Unity Gain Bandwidth PSRR COMP Sink Current COMP Source Current VCOMP Low VCOMP High Slew Rate 9V VCC 18V -40C TJ 125C Note 5 Oscillator Output Frequency IVBIAS = 0mA; VOSC = 0V (Oscillator OFF) 9V VCC 18V, IVBIAS = 0mA; VOSC = 0V 0mA IVBIAS 5mA; VOSC = 0V Rt = 9.47K, Ct = 470pF 4.85 V V mV mV kHz kHz % % Units
Input Bias Current (IFB)
VFB = 2.3V; ICOMP = +500A SINK
SOURCE
August 2006
3
M9999-080206
MIC9131
Parameter Preregulator Input Leakage Current VCC Gate Lockout (VGLO(ON)) VCC Gate Lockout Hysteresis (VGLO) VLINE = 48V VLINE = 180V, VCC = 10V VLINE = 48V 0.1 7.2 700 7.7 500 9 7.5 800
VGLO(ON)
Micrel, Inc.
Condition Min Typ Max 10 Units A V mV V mV mA 1.3 10 10 0.888 50 2.2 +1 1.6 2.5 1.16 4 1.22 140 145 25 VISNS = 5V 0 8 4 12 8 12 6 6 1.28 mA A A V ns A V mV A V A V mV C C ns ns ns
VCC Pre-Regulator Off (VPR(OFF)) VCC Pre-Regulator Hysteresis (VPR) Start-up Current Supply Supply Current, IVCC Enable Input Current Shutdown Supply Current Protection and Control Current Limit Threshold Voltage Current Limit Delay to Output Current Limit Source Current Enable Input Threshold (Turn-on) Enable Input Hysteresis CPWR Input Current CPWR Threshold Soft-Start Current Line UVLO Threshold (Turn-on) Line UVLO Threshold Hysteresis Thermal Shutdown Thermal Shutdown Hysteresis MOSFET Driver Output Minimum On-Time Output Driver Impedance Rise Time Fall Time
Note 1. Note 2. Note 3. Note 4. Note 5.
VLINE = 48V VLINE = 48V VLINE = 48V, VCC = 7.5V, Note 4 Pin 16 (OUT) = OPEN VEN = 0V ,10V; VLINE = 48V VEN = 0V ; VCC = 18V
+0.5V 700 12 1
-10
0.1 0.1
0.772 VISNS = 0V to 5V VISNS = 0V 30 1 -1
0.83 34 40 1.6 150
VCPWR = 5V, 0V VSS = 0V
SINK ; ISINK = 200mA COUT = 500pF COUT = 500pF
SOURCE ; ISOURCE = 200mA
Exceeding the absolute maximum rating may damage the device. The device is not guaranteed to function outside its operating rating. Devices are ESD sensitive. Handling precautions recommended. If a substained DC voltage >150V is applied to the LINE pin, a current-limiting 1.8kresistor should be used in series with the LINE pin. This condition does not apply for transient conditions over 150V. For oscil tions of the internal 5V regulator. See Applications Information for details. -
M9999-080206
4
August 2006
MIC9131
Micrel, Inc.
Typical Characteristics
Oscillator Frequency vs. VCC Voltage
OSC FREQ. VARIATION (%)
0.3
OSC FREQ. VARIATION (%)
2 1.5 1 0.5 0 -0.5 -1 -1.5 -2 -2.5 -40
Oscillator Frequency vs. Temperature
VC C = 10V RB IAS = 560K R = 9.47K t C = 470pF t
2.502
REFERENCE VOLTAGE (V)
Error Amp Reference Voltage vs. V CC Voltage
RB IAS = 560K
0.2 0.1 0.0 -0
FOSC (NOM)=200kHz Rt=9.47K
2.501
-0.2 -0.3 -0.4 -0.5
2.500
Ct=470pF 8 9 10 11 12 13 14 15 16 17 18 VCC (V)
0 40 80 120 TEMPERATURE (C)
160
2.499
8 9 10 11 12 13 14 15 16 17 18 VCC (V)
2.510
REFERENCE VOLTAGE (V)
Error Amp Reference Voltage vs. Temperature
VC C = 10V
1.220 1.215
THRESHOLD (V)
Line UVLO Threshold vs. V CC
1.24
UVLO THRESHOLD (V)
Line UVLO Threshold vs. Temperature
VC C=10V RB IAS=560K
2.505 R B IAS = 560K 2.500 2.495 2.490 2.485 2.480 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
1.210 1.205 1.200 1.195 1.190 1.185 1.180 8 9 10 11 12 13 14 15 16 17 18 VCC (V)
1.23 1.22 1.21 1.2
1.19 1.18 -40 0 40 80 120 TEMPERATURE (C) 160
2.4
QUIESCENT CURRENT (mA)
Quiescent Current vs. V CC Voltage
R = 9.47K t Ct = 470pF
B IAS
1.5
QUIESCENT CURRENT (mA)
Quiescent Current vs. Temperature
QUIESCENT CURRENT (mA)
2.2 2.0 1.8 1.6 1.4 1.2 1.0 8
R
= 560K
1.48 1.46 1.44 1.42 1.4 1.38 1.36 1.34 1.32
VC C = 10V RB IAS = 560K R = 9.47K t Ct = 470pF
10 9 8 7 6 5 4 3 2 1 0
Quiescent Current vs. Frequency
Ct = 470pF
C = 100pF t
300 350 400 450 500 2000
3.5
QUIESCENT CURRENT (mA)
Quiescent Current vs. R BIAS
VC C = 10V R = 9.53K t Ct = 470pf
ISNS to Gate Output Delay vs. R BIAS
90 80 70 DELAY (ns) DELAY (ns) 60 50 40 30 20 10 0 0
200 180 160 140 120
ISNS to Gate Output Delay vs. Overdrive
RB IAS=160K
3
2.5 2
fOSC = 200kHz
1.5 1 0 200 400 600 800 1000 1200 RBIAS (k)
100 80 60 40 20 0
0 200
400 600 800 1000 1200 1400
OVERDRIVE (mV)
August 2006
5
1600 1800
0.5
200 400 600 800 1000 1200 RBIAS (k)
0 50 100 150 200 250
10
12 14 VCC (V)
16
18
1.3 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
GATE DRIVE FREQUENCY (kHz)
RB IAS=560K
RB IAS=360K
M9999-080206
MIC9131
Micrel, Inc.
V BIAS vs. V CC
5.012 5.010 5.008
VBIAS (V)
RB IAS = 560K
5.06 5.04
BIAS VOL AGE (V)
V BIAS Voltage vs. Temperature
V BIAS Load Regulation
5.01 5
VBIAS VOLTAGE (V)
VC C = 10V
5.006 5.004 5.002 5.000 4.998 8 9 10 11 12 13 14 15 16 17 18 VCC (V)
5.02 5
VC C = 10V RB IAS = 560K Rt = 9.47K Ct = 470pF
4.99 4.98 4.97 4.96 4.95 4.94 0 1 2 3 IBIAS (mA) 4 5
4.98 4.96
4.94 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
V CC Turn On/Off Thresholds vs. Temperature
7.8 7.6 THRESHOLD (V) 7.4 7.2 7 6.8 6.6
ISNS Current Limit Threshold vs. V CC Voltage
822.0 821.5 THRESHOLD (mV) 821.0 820.5 820.0 819.5 819.0 818.5 818.0 8 9 10 11 12 13 14 15 16 17 18 VCC (V)
ISNS Current Limit Threshold vs. Temperature
836 834 832 830 828 826 824 822 820 818 816 -40
VC C= 10V RB IAS= 560K
Vcc G LO On
V
RB IAS=560K
CC
=10V
Vcc G LO Of f
6.4 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
THRESHOLD (mV)
RB IAS=560K VC C = 10V
0 40 80 120 TEMPERATURE (C)
160
SHORT CIRCUIT CURRENT (mA)
SINK/SOURCE CURRENT (A)
THRESHOLD VOLTAGE (V)
2 1.95 1.9 1.85 1.8 1.75 1.7 1.65
Enable Threshold vs. V CC
2.5 2
Gate Drive Current vs. V CC
Peak Short Circuit Depletion FET Current vs. Temperature
85 80 75 70 65 60 55 50 45 40 -40 0 40 80 120 TEMPERATURE (C) 160
180V Line
VC C = 0V
S INK
1.5 1 0.5
48V Line
1.6 1.55 1.5 8 9 10 11 12 13 14 15 16 17 18 VCC (V)
SOURCE
0 8 9 10 11 12 13 14 15 16 17 18 VCC (V)
SHORT CIRCUIT CURRENT (mA)
Peak Short Circuit Depletion FET Current vs. V LINE
85 80 70 60 55 125C 50 45 40 0 40 120 80 VLINE (V) CURRENT (mA) 75
16 14 12 10 8 6 4 2 0 0
Depletion FET Current vs. V LINE
-40C
CURRENT (mA)
12 10
Depletion FET Current vs. Low V LINE Voltage
-40C
-40C
25C
25C
8 6 4 2 0 7 7.5 8 8.5 9 VLINE (V)
125C
65 25C
125C
VC C = 0V
160
200
40
VC C=7.5V RB IAS=560K E n =7.5V 120 160 200 80 VLINE (V)
9.5
10
M9999-080206
6
August 2006
MIC9131
Micrel, Inc.
42 ISNS CURRENT (A) 41
ISNS Pin Source Current vs. V CC
RB IAS=560K
ISNS CURRENT (A) 10 12 14 VCC (V) 16 18
41.5 40.5 40 39.5 39 38.5 38 8
45 44 RB IAS=560K 43 VC C=10V 42 41 40 39 38 37
ISNS Pin Source Current vs. Temperature
36 35 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
August 2006
7
M9999-080206
MIC9131
Oscillator Frequency vs. RC Values
47pF
Micrel, Inc.
1000000
RESISTOR VALUE ()
100pF
100000 220pF
470pF 680pF
10000
1000pF
2200pF
1000 10000
Note:
100000 1000000 10000000 FREQUENCY (Hz)
*See applications section for higher switching frequencies
Output switching frequency is 1/4 the oscillator frequency.
Functional Block Diagram
FB
COMP
OSC
SYNC
7
6
4
5
Oscillator
2
SR Latch
Q
Error Amplifier
2.5V
1.2V
2
VCC
16 15
R
OUT
ISNS 14
40A
PWM
PGND
S1 S2
11
AGND
VBIAS EN
9 5V 10
0.82V
RBIAS 3
BIAS REG
1.21V
5V 4A
12
Peak Current Limit
MAXIMUM DUT CYCLE Y
SS
Max. Duty Cycle
CPWR 8
Q R1 R2
Current Limit Selection
VCC 2
S
1-Shot
VCC UVLO
LINE 1
UNDERVOLTAGE LOCKOU T
Thermal Shutdown
13
LINE UVLO
UVLO
Figure 1
M9999-080206
8
August 2006
MIC9131
Micrel, Inc. * Current sensing & overcurrent protection * Slope compensation * Error amplifier High Voltage Start Up Circuit Many conventional Off-Line and Telecom power supplies use an external bias resistor and zener diode to supply the initial start-up voltage for the control IC. The control IC gets its supply voltage from a bias winding once the power supply is running. This method has the disadvantages of extra components (diode and power resistor), continuous power dissipation in the resistor and a large bias capacitor, used to supply the IC until the bias winding takes over. The MIC9131 eliminates these problems by using an internal depletion mode MOSFET as a pre-regulator to provide the start-up bias voltage from the high voltage input of the power supply. This approach eliminates the need for external start up components and reduces the size of the controller's bias supply capacitor. The MOSFET is turned off once the external bias winding takes over, which eliminates power dissipation in the start-up circuit. In some cases, the MIC9131 may be run directly from the input voltage rail, eliminating the need for an external bias winding. Start-up circuit operation is illustrated in Figure 2. VIN is ap-
Functional Description
Micrel's MIC9131 is a high voltage, high speed current mode switching power supply controller. It uses a BiC/DMOS process to achieve a high voltage input, low quiescent current and very fast internal delay times. The MIC9131 is designed to drive an external low side N-channel MOSFET, which makes it suitable for controlling Boost, Flyback and Forward converter topologies. The high voltage startup pin eliminates the requirement for an external start up circuit. This makes it ideal for use with Telecom converters. A block diagram of the MIC9131 is shown in Figure 1. The description of the controller is divided into 6 basic functions: * Power and bias circuitry * High voltage start-up circuit * VCC and Bias supplies * Enable and Undervoltage monitoring circuits * VCC and VIN UVLO * Enable * Oscillator and sync circuitry * Soft-start and soft-start reset circuits * MOSFET gate drive circuits * Control loop operation
Transformer Bias Winding
MIC9131
VCC 2
Internal Circuitry
1.21V
VIN
Line 1
180V DEPLETION FET
VCC UVLO
THERMAL SHUTDOWN
VPR(OFF) Depletion FET Pre-Regultor turn-off threshold VPR Depletion FET turn-on threshold VGLO(ON) VCC gate lockout turn on threshold
VGLO VCC Gate Lockout Hysteresis
VCC voltage when powered from VLINE
Figure 2
August 2006
9
M9999-080206
MIC9131 plied and the depletion FET, which is normally enabled allows current from VIN to charge the VCC bias capacitor. Once the VCC voltage reaches the VCC enable threshold, VGLO(ON) , the gate drive is enabled and the MIC9131 starts switching. Vcc continues to increase until the Pre-Regulator turn-off threshold, (VPR(OFF)), is reached and the depletion FET is turned off. The Vcc voltage decreases as energy from the bias capacitor is used to supply the controller. The depletion FET is turned back on when the pre-regulator turn-on threshold is reached. A bias winding derived supply voltage, set higher than the FET turn-off threshold, VPR(OFF), raises the VCC voltage over the threshold and prevents the FET from turning on. In certain designs the MIC9131 may be powered directly from the Line voltage, eliminating the need for an extra transformer bias winding. When operating in this fashion the designer must insure the power dissipation in the IC does not cause the die temperature to exceed the 125C maximum. Power dissipation is calculated by: PDISS = ( V - VCC ) x IVCC IN Where :
Micrel, Inc. VPR hysteresis). The bias regulator in the MIC9131 buffers the internal circuits from VCC variations. The pre-regulator FET is protected by a thermal shutdown circuit, which turns the MOSFET off if its temperature exceeds approximately 150C. When operating at input voltages greater than 150V, a fast input voltage risetime during turn-on (which may occur during a hot plug operation)may cause a high peak current to flow through the depletion FET, damaging the MIC9131. A 1.8k resistor in series between the input voltage and the Line pin (pin 1) is recommended when operating at input voltages greater than 150V. This resistor limits the maximum peak current to 100mA (at 180VIN) and protects the part. The depletion mode MOSFET contains an internal parasitic diode. The VIN pin voltage must be greater than the VCC voltage or the VCC voltage will be clamped to a diode drop greater than the VIN voltage. Excessive power dissipation in the parasitic diode will destroy the IC. VCC and Bias Supplies
VIN is the line input voltage VCC is the average VCC voltage (typically 8.5V) IVCC is the total current drawn by the IC IVCC is the sum of the operating current of the MIC9131 at a given frequency and the average current required to drive the external switching MOSFET. A plot of typical operating current vs. frequency is given in Figure 3. The average MOSFET gate drive current is calculated in the "MOSFET GATE DRIVE" section of this specification.
10 9 8 7 6 5 4 3 2 1 0 QUIESCENT CURRENT (mA)
Quiescent Current vs. Frequency
Ct = 470pF
C = 100pF t
150 200 250 300 350 400 450 500
GATE DRIVE FREQUENCY (kHz)
Figure 3 The die junction temperature is calculated by TJ = TA + PDISS x JA Where: TJ is the die junction temperature TA is the ambient temperature of the circuit JA is the junction to ambient thermal resistance of the MIC9131 (listed in the operating ratings section of the specification. When powered directly from the Line voltage, the VCC voltage will vary between the upper and lower pre-regulator thresholds. The amplitude of the output gate drive voltage will vary with the VCC voltage. This should not be a problem for most topologies since the variation is small (equal to the
M9999-080206
The power for the controller and gate drive circuitry is supplied through the VCC pin. The gate drive current is returned to ground through the power ground pin (PGND). The rest of the supply current is returned to ground through the analog ground pin (AGND). The two ground pins must be connected together through the PCB ground plane. High frequency decoupling is provided at the VCC pin to supply the gate drive's peak current requirements. Turn-on of the external MOSFET causes a voltage glitch on the VCC pin. If the glitch is excessive, this disruption can appear as noise or jitter in the oscillator circuit or the gate drive waveform. The decoupling capacitor must be able to supply the MOSFET gate with the charge required to turn it on. A 0.1F ceramic capacitor is usually sufficient for most MOSFETs. Larger FETs, with a higher gate charge requirement may require a 0.22F ceramic capacitor or a ceramic capacitor paralleled with a 2.2F tantalum or 4.7uF aluminum electrolytic. It is recommend that if VLINE is greater than 150V DC than the maximum capacitor recommended on VCC is 2.2F.The capacitor must be located next to the VCC pin of the MIC9131. The ground end of the capacitor should be connected to the ground plane, making a low impedance connection to the power ground pin (pin 15). The internal bias regulator block provides several internal and external bias voltages. Referring to Figure 1, a 2.5V reference is used for the internal error amplifier, a 0.82V bias is used by the current limit comparator and a 1.21V reference is used by the Line UVLO circuit. An external 5V bias voltage (VBIAS) powers the oscillator circuit and may be used as a reference voltage for other external components. The VBIAS pin requires a minimum 0.1F capacitor to ground for decoupling. Enable and Undervoltage Monitoring Circuits The two undervoltage lockout circuits in the MIC9131 are shown in Figure 4. One monitors the VCC voltage and the other monitors the input line voltage. These signals are OR'd together and either one can disable the gate drive pin and discharge the voltage on the soft start capacitor. 10 August 2006
0 50 100
MIC9131
5V
Micrel, Inc.
MIC9131
4A
12
SS
SET S Q
VCC
2
R 1.21V VCC UVLO RESET
/Q
UVLO VIN R1 R2 AGND
11
16
OUT
UVLO
13
LINE UVLO
15
PGND
Figure 4: UVLO and Soft Start Circuits VCC Undervoltage Lockout The VCC voltage is internally divided down and compared to a 1.21V internal bandgap reference. As VCC rises above the turn-on threshold, it disables the VCC undervoltage lockout circuit. Once above the turn-on threshold, hysteresis prevents the lockout circuit from disabling the IC until the VCC voltage falls below the lower threshold. Line Undervoltage Circuit (UVLO) The line voltage is monitored by an external resistor divider and fed into the negative input of the line UVLO comparator. As the comparator trip point is exceeded, the line UVLO circuit is disabled. Hysteresis built into the comparator prevents the circuit from toggling on an off in the presence of noise or a high input line impedance. The line voltage turn-on trip point is: R2 VLINE_ON = VTHRESHOLD x R1+ R2 where: VTHRESHOLD is the voltage level of the internal comparator reference, typically 1.21V. The line hysteresis is equal to: R1 + R2 VHYSTERESIS = V HYST x R2 where: VHYST is the internal hysteresis level, typically 75mV. VHYSTERESIS is the hysteresis of the line input voltage The MIC9131 will be disabled when the line voltage drops back down to: V LINE_OFF = V LINE_ON - VHYTERESIS= Enable A low level on the enable pin turns off all the functions of the MIC9131 and places it in a low quiescent current state. The output driver is in a low state. When the enable pin is pulled high, the MIC9131 goes through its normal start up sequence including undervoltage lock out and soft start. When not used, the pin should be connected to VCC. Oscillator Block An external resistor and capacitor set the oscillator frequency. The MIC9131 contains an internal divide-by-four circuit that limits the maximum duty cycle at the gate drive to 75%. The oscillator frequency of the MIC9131 is four times the output switching frequency. Oscillator Pin The operation of the oscillator is shown in Figure 5. The voltage waveform at the OSC pin is a sawtooth whose amplitude increases as capacitor Cosc is charged up through ROSC from the 5VBIAS. When the OSC pin voltage reaches the internal comparator upper threshold, COSC is quickly discharged to zero volts by an internal MOSFET. After a brief delay, typically 75ns, the internal MOSFET is turned off and the COSC charges, repeating the cycle. Figure 5 show the relationship between the oscillator and gate drive waveforms. The delays in the IC force the duty cycle of the gate drive signal to be slightly less than 75% duty cycle. For VBIAS = 5V and a peak oscillator waveform voltage of 3V, the design equations simplify to: Charging t CHARGE = 0. 92 x R t x Ct Discharging tDISCHARGE 40 x C t
(VTHRESHOLD - VHYST )x R1R2R2 +
August 2006 11
M9999-080206
MIC9131 TP_OSCILLATOR = t CHARGE + t DISCHARGE + t DELAY Where t DELAY = 75ns fS _ OSCILLATOR = fS _ OUTPU = 1 TP _ OSCILLATOR
2N3904
ROSC 1.6k COSC 33pF
Micrel, Inc.
VCC
2
5 9
1F
SYNC VBIAS
4.7F
OSC
4
3V
1 x fS _ OSCILLATOR 4
4.7F
The timing capacitor, COSC, should be an NPO ceramic or a temperature stable film capacitor. Care must be taken when using capacitor values less than 47pF. The high impedance of a small value capacitor makes the OSC pin more susceptible to switching noise. Also, the input capacitance of the OSC pin and the stray capacitance of the board will have a noticeable effect on the oscillator frequency.
SYNC VBIAS ROSC COSC 75ns 1-shot
11 5 9
AGND
11
75ns 1-shot
Figure 5a Oscillator Synchronization The switching frequency of the MIC9131 can be synchronized to an external oscillator or frequency source. Figure 6 shows the relationship between the sync input, oscillator waveform and gate drive output. The external frequency should be set at least 15% greater than the free running oscillator frequency to account for tolerances in the oscillator circuit and external components. The positive edge of the sync signal resets the oscillator. The sync pulse frequency, like the oscillator, is four times the gate drive frequency. When an external sync signal is applied, the peak amplitude of the oscillator signal (pin 4) is less than when it is free running because the oscillator signal is terminated before it reaches its 3V (typical) amplitude. When not used, the sync pin should be connected to ground to prevent noise from erroneously resetting the oscillator.
Sync Input (pin 5) Gate Drive (pin 16) Oscillator Waveform (pin 4)
3V
OSC
4
AGND
VOSC
Gate Drive (pin 16) tON tPERIOD
Figure 5 Higher Switching Frequencies The MIC9131 is capable of very high switching frequencies. One of the limitations on the maximum frequency is the current capability of the 5V regulator supplying the oscillator and VBIAS. By powering VBIAS with an external source, e.g. linear regulator much higher switching frequencies can be achieved. A simple way of using an external current source is to set an NPN as an emitter follower. Figure 5b shows the MIC9131 oscillator frequency set to 4MHz using an external NPN. The emitter followerj circuit allows the current to be supplied by VCC while the voltage is regulated to a diode drop below VBIAS. This configuration is quite stable over temperature and voltage variations.
TIME (500ns/div)
Figure 6. Sync Waveform Soft Start Circuit The soft start is programmed by a capacitor on the soft start pin. A 4A current source charges up the capacitor. At power up, the SS pin is discharged. Once the UVLO and enable functions release the soft start circuit, the voltage of the capacitor increases. The active voltage range of the soft start pin is from typically from 0.9V to 1.7V. The internal current source increases the voltage on the soft start capacitor to approximately 4V. The soft start pin and the current sense voltage are connected to a comparator in the MIC9131. The voltage from the soft start pin effectively limits the peak current through the current sense resistor by prematurely terminating the on-time of the gate drive output. Referring to Figure 1, with the soft start voltage low, the duty cycle of the output is at a minimum. As the soft start voltage increases, the duty
M9999-080206
12
August 2006
MIC9131 cycle of the gate drive output increases until the error amplifier takes control of the duty cycle. The soft start capacitor is discharged by an internal MOSFET in the MIC9131. The soft start circuit is activated by the following events: 1. Line undervoltage pin less than the 1.21V threshold 2. VCC becomes less than the pre-regulator voltage turn off threshold. 3. The current limit comparator threshold is exceeded. This can be disabled with a low level on the CPWR pin. 4. A low level on the enable pin. Calculating the soft capacitor depends on many parameters such as the current limit of the circuit input voltage, output power and output loading. A starting value of capacitor should be chosen and the value can be adjusted later in the design. Recommended starting values of soft start capacitance is typically 10nF to 100nF. Values below 1nF may be ineffective in slowing the output voltage turn on time. CPWR Current Limit Selection This pin controls whether the soft start circuit is reset if the voltage on the Isns pin exceeds the overcurrent threshold. When the CPWR pin is high, an overcurrent condition at the ISNS pin will terminate the on-time of the gate drive pulse and discharge the soft start capacitor to 0V. This delay in start up contributes to a reduction in the average output current during an overcurrent or short circuit condition. A smaller MOSFET may be used since the power dissipation in the MOSFET is minimized under short circuit or overcurrent conditions. If the CPWR pin is low an overcurrent or short circuit conditions will not trip the soft start circuit. The pulse-by-pulse current limit, inherent in current mode control, provides a "brick wall" or constant current limit. With the power supply operating in this mode, a smaller soft start capacitor can be used to increase the turn on speed of the supply. If the CPWR in is held low during the initial turn on at power up and then raised high, the power supply can maximize the turn-on time at start up and still provide a high level of overcurrent and short circuit protection. The circuit shown in Figure 7 performs this function.
MIC9131 VREF
Micrel, Inc. of the output is dependent on the IC supply voltage and the gate charge required to turn the MOSFET on and off. A resistor placed in series with the gate drive output attenuates ringing in the etch connection between the MIC9131 and the MOSFET. Figure 8 shows a single resistor in series between the driver output and the gate of the MOSFET. The zener value should be greater than the gate drive voltage to prevent excessive power dissipation, but less than the maximum gate to source voltage rating.
Gate Drive Output
GND
Figure 8 The circuitry shown in figure 9 allow different rise and fall times. R1 and the input capacitance of the MOSFET determine the rise-time of the gate voltage and therefore the turn-on time of the MOSFET. The diode, D1 is reversed biased, which removes R2 from the circuit. At turn-off, D1 is forward biased and the parallel combination of R1 and R2 controls the turn-off time of the MOSFET. The turn on-time is slower, which reduces switching noise and ringing during turn-on. The turn-off time is faster, which minimizes switching losses during turn-off and improves efficiency. If the turn-on time is to be faster than the turn-off time, the diode should be reversed.
R2 D1 R1
Gate Drive Output
GND
Figure 9 A gate drive transformer is used where an increase in drive voltage, isolation and/or voltage level shifting are required. Gate drive transformers can have multiple windings and drive multiple MOSFETs, including MOSFETs that require a drive signal 180C out of phase with the ICs drive signal. Figure 10 shows a gate drive transformer circuit. The capacitor, C1 removes DC from the drive circuit and prevents transformer saturation. R1 provides damping to eliminate ringing in the circuit. R1 is usually in the 5 to 20 range, depending on the amount of damping necessary. D1 and D2 form a clamp circuit, which prevents the voltage from exceeding the VGMAX level. If the gate drive is well damped, the diodes may be removed R2 is used to allow the transformer to reset properly.
D1
R1 CPWR
C1
AGND
Figure 7 MOSFET Gate Drive Output The MIC9131 has the capability to directly drive the gate of a MOSFET. The output driver consists of a complimentary P-channel and N-channel pair. The typical switching time
August 2006
13
M9999-080206
MIC9131
C1
Micrel, Inc. Current Sense Circuit The current sense input of the MIC9131 has three unique features, which are advantageous in a high speed, high efficiency power supply. 1. The overcurrent threshold is nominally 0.82V instead of the typical 1.0V found in most switching control ICs. 2. The current sense pin sources a nominal 40A of cur rent out of the pin. This is used to raise the current limit threshold of the pin, which allows a smaller current sense resistor to be used. This improves the efficiency of the power supply, especially in lower current applications. 3. The delay from the current sense input to the output is typically 50ns. The current limit threshold of the ISNS pin was set at 0.82V, allowing the use of a smaller current sense resistor. A stable, bandgap derived 40A current is sourced from the ISNS pin. A voltage drop across a series resistor placed between the pin and the current sense resistor level increases the current sense signal at the ISNS pin. This allows the use of a smaller current sense resistor if the full 0.82V peak to peak current signal is not required. Decreasing the value of the current sense resistor decreases the power dissipation in the resistor, which improves the efficiency of the power supply. The delay between the input of the overcurrent comparator and the output gate drive is nominally 50ns. This very fast response time allows the MIC9131 to operate at higher frequencies and still have adequate overcurrent protection. The operation of the current sense input is as follows. The sensed current in the power supply is converted to a voltage by a resistor or current sense transformer. Referring to Figure 1, this voltage is compared to the output of the error amplifier, which sets the duty cycle of the gate drive output. The current signal is also connected to an Imax comparator. Comparing the current sense signal to the reference voltage sets a maximum current limit. If the maximum amplitude of the current sense signal exceeds the reference, the comparator terminates the gate drive output pulse. It aslo discharges the soft start capacitor when the CPWR pin is high. Leading Edge Current Spike The current signal in a power circuit will often have a leading edge spike caused by leakage inductance, parasitic inductance and capacitance, diode reverse recovery effects and snubbers. These spikes can cause premature termination of the switching cycle if they are not eliminated. A resistor may be added in series between the current sense resistor and the Isns input. The input and board trace capacitance of the ISNS pin (pin 14) is approximately 25pF. A 1k resistor is a good choice, since it attenuates most of the ripple without distorting the current sense waveform. It has a minimal effect on level, offsetting the current sense signal by only 40mV. A typical rule of thumb is the bandwidth of the RC filter should be at least 6 times the switching frequency. This avoids distorting the current sense waveform and adding excessive delays in the current loop that will interfering with overcurrent protection. For a 100kHz switcher, the maximum 14 August 2006
T1
R1 D2 R2 D1
Gate Drive Output GND 1:N
Figure 10 The gate impedance of a MOSFET is capacitive and the power required to drive the gate is proportional to the charge required to turn on the MOSFET, the peak gate voltage and the switching frequency. Assuming the total gate charge for turn on and turn off is equal, the power used to switch the MOSFET on and off is: PDRIVE = QG x VGS x fS where: QG is the total gate charge at VGS VGS is the gate to source voltage of the MOSFET usually equal to VCC fS is the output switching frequency The power required to drive the MOSFET is dissipated in the drive circuitry of the MIC9131. This power must not cause the die temperature to exceed the maximum rated junction temperature of 125C. MOSFET Driver IC's are used when the drive requirement for the MOSFETs is greater than the capability of the MIC9131 gate drive output. While the peak current of the MIC9131 gate drive is typically 1.2A at VIN =12V, a gate driver ICs will sink or source between 1.2A and 12A of peak current. The higher peak current allows faster rise and fall times for larger MOSFETs. The drive requirements for selecting a MOSFET driver are determined using the following equation: Q IPK = 2 x G t where: QG is the total gate charge required to turn on the MOSFET at a specified ID, VG and VDS. This information is usually given in the MOSFET specification sheet. t is the gate voltage transition time (risetime or fall time) IPK is the peak current requirement of the MOSFET driver IC. For example, if a MOSFET is chosen with a QG of 60nC and it is desired to have a 50nS gate to source risetime/falltime, the peak current requirement of the MOSFET driver is: 2 x 60nC IPK = = 2. 4A 50ns A driver such as the MIC4424 will meet this requirement. For more information on choosing a MOSFET driver, see the Micrel application note AN-24, "Designing with Low Side MOSFET Drivers."
M9999-080206
MIC9131 series resistance is 10K, for a 500kHz switcher, the maximum series resistance is 2K. Sensing Current with a Resistor The fast transition times of the current signal prohibit the use of inductive resistors. Standard wire wound power resistors will not work. Carbon composition or metal film resistors or low inductance power resistors may be used. The overcurrent range of the power supply and component tolerances must be considered when selecting the current sense resistor value. The power supply specification may call for an overcurrent limit, which must be accounted for when selecting the current sense resistor value. The relationship between the peak primary current and the current sense resistor is: VISNS = IP x RISENSE + IISNS x R f where: Ip is the current in the sense resistor RISENSE is the current sense resistance IISNS is the current sourced from the ISNS pin (40A) Rf is the series resistor between the ISNS pin and the current sense resistor. The current sense resistor must not be too small or the current sense signal will be susceptible to noise. If noise is a problem, the current signal level should be increased. An example is illustrated below. The maximum peak current, IPMAX= 1A at 120% overcurrent and minimum input voltage The maximum rms current, IRMS=0.65A The desired current sense signal amplitude is 500mV at 1A output current. The current sense resistor value and power dissipation is: V 0.5 RSENSE = SENSE = = 0.5 ISENSE 1 PDISS = IRMS2 x RSENSE = 0. 65 2 x 0 .5 = 0.21W A 0.5 ohm, non inductive resistor with at least a 1/2W rating should be selected. The series resistor is calculated to allow the 500mV-peak signal to reach 0.82V. Rf VISNS - ( IP x RISENSE ) IISNS = 0. 82 - (1x 0 .5) = 10.25k 40 A
Micrel, Inc. Sensing Current with a Current Sense Transformer At higher power levels, the power dissipation in a current sense resistor is excessive. A current sense transformer can be used to sense the current while minimizing power dissipation. See Figure 11. The schematic shows the circuitry necessary when using a current sense transformer. The resistor, R1, provides a path to reset the current sense transformer. The resistor, R2, converts the scaled down current to a voltage, which is sent to the ISNS pin.
VIN
ISNS (pin 14) MIC9131 OUT (pin 16)
Rf R2 R1
Current Sense Transformer IPRI
Figure 11 The voltage at the ISNS pin is calculated by: I VISNS = P x R2 + IISNS x R f N where: IP is the current in the primary of the current sense transformer R2 is the current sense resistance at the secondary of the current sense transformer N is the turns ratio of the current sense transformer (N=Nsec/Npri) IISNS is the current sourced from the ISNS pin (40A) Rf is the series resistor between the ISNS pin and the current sense resistor. Current Transformer Example: The maximum peak current, IPMAX = 5A at 120% overcurrent and minimum input voltage The maximum rms current, IRMS = 3.25A The full 0.82V peak signal a the ISNS input can be used since very little power is dissipation in the secondary side sense resistor. The maximum peak to peak voltage at the sense pin (pin 14) is 0.82V at the 5A maximum output current. The current sense resistor value and power dissipation is: V x N 0. 82 x 100 = 16.4 R2 = SENSE = IP 5
I 3. 25 PDISS = PRMS x R2 = x 16.4 = 17 .4 mW 100 N
2 2
The next lower value of 10k is selected. The bandwidth of the 10K resistor and the 25pF input capacitance is calculated. The resistor value must be lowered if the bandwidth is too low for the switching frequency. 1 BW = = 630kHz 2 x x 10k x 25 pF The maximum switching frequency of this power supply should be approximately six times less than the BW to prevent current waveform distortion and excessive delays in the current loop. This limits the switching frequency to the range of 100kHz.
A 16.2 ohm, 1%, non inductive resistor with at least a 50mW rating should be selected. A good choice would be an 0805 size metal film or a 1/8 watt leaded metal film resistor. A series resistor between the current sense transformer and the Isns input is not necessary unless it is used for low pass filtering. 15
M9999-080206
August 2006
MIC9131 If the current sense transformer were not used, the sense resistor would dissipate 1.7 watts. V 0. 82 RSENSE = SENSE = = 0.164 ISENSE 5 PDISS = IRMS2 x RSENSE = 3. 25 2 x 0 .164 = 1. 7 W Slope Compensation Power supplies using peak current mode control techniques require slope compensation when they are operating in continuous mode and have a duty cycle greater than 50%. Without slope compensation, the duty cycle of the power supply will alternate wide and narrow pulses commonly referred to as subharmonic oscillations. Even though the MIC9131 operates below a 50% duty cycle, slope compensation adds the benefits of improved transient response and greater noise immunity in the current sense loop (especially when the current ramp is shallow). Slope compensation can be implemented by adding an optimum 1/2 of the inductor current downslope, reflected back to the current sense input. In real world applications, 2/3 of the inductor current downslope is used to allow for component tolerances. Slope compensation at the ISNS input may be implemented by using a resistor and capacitor as shown in Figure 12. The rectangular waveshape of the gate drive output is integrated by the resistor/capacitor filter, which results in a ramp used for the slope compensation signal. When the gate drive and the current signal at the sense resistor goes low, the capacitor is discharged to 0V.
Gate Drive (pin 16) MIC9131 ISNS (pin 14) R2 R1 C1
Micrel, Inc. secondary winding inductance for the flyback topology) M2 is the inductor current downslope For a boost topology, the inductor downslope is: di VOUT - VIN + VD M2 = = dt L In a transformer isolated topology, the downslope must be reflected back to the primary by the turns ratio of the transformer. The reflected downslope is: Ns M2REFLECTED = M2 x Np where : Ns/Np is the turns ratio of the secondary winding to the primary winding. M2REFLECTED is the inductor curent downslope reflected to the secondary side of the current sense transformer. The reflected downslope is multiplied by the current sense resistor to obtain the downslope at the current sense input pin (ISNS). ISNS _ SLOPE = M2REFLECTED x RS where Rs is the value of the current sense resistor. The required downslope of the compensation ramp at the ISNS input is: M3 = ISNS _ SLOPE x 0.67 R1 is know if a value for the resistor between the current sense resistor and the Isns pin, has already been selected. If not chose a value of 1k, which will minimize any offset and signal degradation at the ISNS pin. Select a value of C1 to minimize signal degradation from the cutoff frequency of R1/C1. The bandwidth should be at least six times the switching frequency. 1 C1 = 2 x x fS x R1 where: fS is the switching frequency of the power supply (not the oscillator frequency) The slope of the generated compensation ramp is: R1 1 M3 = VGATE_DRIVE x x R2 + R1 R2 x C1 Solving for R2 and assuming R2 is much greater than R1. R2 = VGATE _ DRIVE x R1 M3 x C1
RSENSE
Figure 12 The procedure outlined below demonstrates how to calculate the component values. Compute the inductor current downslope as seen at the current sense input. For a flyback, buck or forward mode topology the inductor downslope is equal to: di VO + VD M2 = = dt L where : VO is the output voltage VD is the forward voltage drop of the rectifier diode L is the inductance of the output inductor (or the
where: VGATE_DRIVE is the amplitude of the gate drive waveform
M9999-080206
16
August 2006
MIC9131 Error Amplifier The error amplifier is part of the voltage control loop of the power supply. The FB pin is the inverting input to the error amplifier. The non-inverting input is internally connected to a 2.5V reference. The output of the error amplifier, COMP, is connected to the PWM comparator. The error amplifier
Micrel, Inc. provides the reference to limit and control the peak current of the power supply. There is a 1.2V level shift between the output of the error amplifier and the PWM comparator. This allows the output of the error amplifier to operate in a linear region and prevents loading on the COMP pin from interfering with proper control of the current signal.
August 2006
17
M9999-080206
MIC9131
Micrel, Inc.
Package Information
16-Pin SOP (M)
16-Pin QSOP (QS)
M9999-080206
18
August 2006
MIC9131
Micrel, Inc.
MICREL INC.
TEL + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com
2180 FORTUNE DRIVE
SAN JOSE, CA 95131
USA
This information furnished by Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Pr Micrel for any damages resulting from such use or sale. (c) 2001 Micrel Incorporated
August 2006
19
M9999-080206

▲Up To Search▲

Price & Availability of MIC913106

	To Download MIC913106 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .